Wide dynamic range transimpedance amplifier with a controlled low frequency cutoff at high optical power

ABSTRACT

A wide dynamic range transimpedance amplifier with a low cut off frequency at high optical power. An automatic transimpedance gain control and DC cancellation control feedback circuit includes variable impedance circuitry. An emitter terminal of a first pnp transistor is connected to the input of the transimpedance amplifier. The impedance seen at the emitter terminal changes according to the average value of the input current. Open loop gain of the feedback loop including the first pnp transistor is not dependent on the average input current as the input current increases. A base terminal of the first pnp transistor is connected to a base terminal of second pnp transistor. Emitter size of the second pnp transistor is some factor N smaller than emitter size of the first pnp transistor. N can be configured to adjust the gain control and low corner frequency variation with input power.

RELATED APPLICATIONS

[0001] The present application is a continuation-in-part of U.S. patentapplication Ser. No. 10/371,847, filed Feb. 21, 2003, and entitled “AWide Range Transimpedance Amplifier With A Controlled Low FrequencyCutoff At High Optical Power, which is hereby incorporated by reference.That application claims the benefit of U.S. Provisional Application No.60/429,129, filed Nov. 26, 2002 and entitled “Circuit for Wide DynamicRange Transimpedance Amplifier,” which is hereby incorporated byreference.

BACKGROUND OF THE INVENTION

[0002] 1. The Field of the Invention

[0003] The present invention relates to a wide dynamic rangetransimpedance amplifier. More particularly, the present inventionrelates to a wide dynamic range transimpedance amplifier with acontrolled low cutoff frequency as optical power received at thetransimpedance amplifier increases.

[0004] 2. The Relevant Technology

[0005] Fiber optic networks use light signals to transmit data over anetwork. Although light signals are used to carry data, the lightsignals are typically converted into electrical signals in order toextract and process the data. The conversion of an optical signal intoan electrical signal is often achieved utilizing a fiber optic receiver.A fiber optic receiver converts the optical signal received over theoptical fiber into an electrical signal, amplifies the electricalsignal, and converts the electrical signal into an electrical digitaldata stream.

[0006] The fiber optic receiver usually includes a photodiode thatdetects the light signal and converts the light signal into anelectrical signal or current. A transimpedance amplifier amplifies thesignal from the photodiode into a relatively large amplitude electricalsignal. The amplified electrical signal is then converted into a digitaldata stream.

[0007] The optical signals that are converted into electrical signals bythe fiber optic receiver, however, can vary significantly in bothamplitude and power. The power of the optical signal is often related,for example, to the length (hence loss) of the optical fiber over whichthe optical signal was transmitted, the laser source power, theefficiency of the photodiode, etc. These and other factors result inoptical signals whose incident power at the transimpedance amplifier canvary significantly.

[0008] Fiber optic receivers are only able to successfully receive andamplify optical signals that fall within a particular power range. Inorder for a fiber optic receiver to accommodate a wide range of opticalsignals, the fiber optic receiver and in particular, the transimpedanceamplifier, should be able to detect and amplify very low levels ofoptical power as well as high levels of optical power. The range ofsignals that can be successfully amplified is therefore effectivelylimited by the incident optical power because the fiber optic receiverdistorts or clamps signals whose optical power is too large and cannotrecognize signals whose optical power is too low.

[0009] One problem with current transimpedance amplifiers is thatextending the ability of the transimpedance amplifier to amplify signalswith more optical power usually diminishes the ability of thetransimpedance amplifier to amplify signals with low optical power. Inother words, the maximum optical input power that can be accepted by thetransimpedance amplifier while meeting signal integrity and bit errorrate specifications is usually specified as the input optical overload.The minimum input power is specified as optical sensitivity. Thetransimpedance amplifier should be designed to maximize the opticaloverload and minimize the optical sensitivity. In most of the commercialor published transimpedance amplifiers, there is a direct tradeoffbetween the circuit optical (or current) sensitivity (or equivalentinput current noise) and the optical (or current) overload. Somesolutions to this problem, such as utilizing clamping circuitry orvoltage regulators to assist in the amplification of optical signalswith relatively large optical power, add both cost and complexity to thetransimpedance amplifier of the fiber optical receiver. Without the aidof additional circuitry, the range of signals that can be successfullyamplified is somewhat limited because the circuitry used for automaticgain control and DC cancellation introduces unwanted gain into thetransimpedance amplifiers DC cancellation feedback loop at large opticalpower.

[0010] The unwanted gain also has an adverse effect on the low frequencycutoff at higher optical powers. In other words, transimpedanceamplifiers do not function at certain frequencies because the lowfrequency cutoff has been increased. The low frequency cutoff for thesetypes of transimpedance amplifiers is related to the transconductance ofthe circuitry used for automatic gain control and DC cancellation. Thus,as the current of the input signal increases, the low frequency cutoffof the transimpedance amplifier is adversely affected.

BRIEF SUMMARY OF THE INVENTION

[0011] These and other limitations are overcome by the presentinvention, which relates to a wide range dynamic transimpedanceamplifier. In the present invention, the wide dynamic range of thetransimpedance amplifier is accomplished in a manner where the gain inoptical overload is not completely offset by a loss of opticalsensitivity. In addition, the low cutoff frequency does not increaselinearly but approaches an upper limit or is controlled as the inputcurrent to the transimpedance amplifier increases. This permits, in oneembodiment, the transimpedance amplifier to be utilized with legacysystems that may operate at lower frequencies. The low cutoff frequencyis controlled as the optical power increases.

[0012] In one embodiment, a transimpedance amplifier includes feedbackcircuitry that provides both automatic gain control, AC attenuation, DCshunting, and a low cutoff frequency at higher optical input powers. Apnp transistor is used in the DC cancellation feedback circuitry suchthat the emitter impedance of the pnp transistor is controlled, via afeedback loop, by the average photodiode current. The emitter is alsoconnected with the input of the transimpedance amplifier.

[0013] As the photodiode current increases in response to increasedoptical power, the emitter impedance of the pnp transistor, which isconnected with the input current or signal, decreases. However, unlike acommon emitter npn transistor whose collector is connected to the inputof the transimpedance amplifier, the pnp transistor does not introducesignificant additional gain into the feedback loop as the input signalamplitude increases, thereby keeping the low-cutoff frequencysubstantially unchanged.

[0014] An npn transistor can also be used, for example, when thetransimpedance amplifier is connected to the emitter of the npntransistor. Also, the npn is used for in embodiments where a photodiodehas its cathode connected to the input of the transimpedance amplifier.

[0015] Automatic gain control is achieved because the AC component ofthe photodiode current is increasingly shunted to ground by the pnptransistor as the average photodiode current increases. The AC componentis attenuated at higher amplitudes. As the average photodiode currentdecreases, the emitter impedance of the pnp transistor decreases andenables low power signals to be passed with little or no attenuationinto the main amplifier. This ensures that the optical sensitivity ofthe transimpedance amplifier is not traded for optical overload. Inanother example, a shunt feedback transimpedance amplifier also includesfeedback circuitry to provide both automatic gain control, ACattenuation, and DC cancellation.

[0016] The variable impedance of the feedback circuitry can be achievedusing a pnp transistor, an npn transistor, field effect transistors, andthe like. In one embodiment, the emitter of an npn transistor isconnected with an emitter of a pnp transistor such that current from thephotodiode can either be sourced or sunk. Photodiodes that amplify theinput current or signal can be accommodated by optimizing, in oneexample, the pnp transistor to trigger earlier.

[0017] In some embodiments of the invention, a transimpedance amplifierincludes a first pnp transistor and a second pnp transistor. The emitterterminal of the first pnp transistor is connected to the input of thetransimpedance amplifier. The base terminal of the first pnp transistoris connected to the base terminal of the second pnp transistor. Theemitter size of the second pnp transistor is some factor N smaller thanthe emitter size of the first pnp transistor.

[0018] Additional features and advantages of the invention will be setforth in the description which follows, and in part will be obvious fromthe description, or may be learned by the practice of the invention. Thefeatures and advantages of the invention may be realized and obtained bymeans of the instruments and combinations particularly pointed out inthe appended claims. These and other features of the present inventionwill become more fully apparent from the following description andappended claims, or may be learned by the practice of the invention asset forth hereinafter.

BRIEF DESCRIPTION OF THE DRAWINGS

[0019] To further clarify the above and other advantages and features ofthe present invention, a more particular description of the inventionwill be rendered by reference to specific embodiments thereof which areillustrated in the appended drawings. It is appreciated that thesedrawings depict only typical embodiments of the invention and aretherefore not to be considered limiting of its scope. The invention willbe described and explained with additional specificity and detailthrough the use of the accompanying drawings in which:

[0020]FIG. 1 illustrates an exemplary environment for implementingembodiments of the present invention;

[0021]FIG. 2 is a block diagram of a transimpedance amplifier thatprovides both automatic gain control and DC cancellation;

[0022]FIG. 3 illustrates one embodiment of the present invention in acommon base configuration with a variable impedance formed using a pnptransistor;

[0023]FIG. 4 illustrates an embodiment of feedback circuitry in atransimpedance amplifier where the variable impedance includes both annpn transistor and a pnp transistor, thereby enabling the variableimpedance to either source or sink the current from the photodiode;

[0024]FIG. 5 depicts a shunt feedback transimpedance amplifier withautomatic gain control and DC cancellation circuitry;

[0025]FIG. 6 illustrates another embodiment of feedback circuitry in atransimpedance amplifier using field effect transistors;

[0026]FIG. 7 plots the transimpedance of a transimpedance amplifierversus the average photodiode current;

[0027]FIG. 8 plots the low cutoff frequency of a transimpedanceamplifier versus the average current of the photodiode;

[0028]FIG. 9 illustrates another embodiment of a transimpedanceamplifier with automatic gain control and DC cancellation circuitry;

[0029]FIG. 10 is an example plot of the gain of a transimpedanceamplifier versus frequency;

[0030]FIG. 11 is an example of a number of plots of transimpedanceamplifier transimpedance gain versus photodiode average current; and

[0031]FIG. 12 is an example of a number of plots of transimpedanceamplifier low frequency cut-off versus photodiode average current.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

[0032] The present invention relates to a wide dynamic rangetransimpedance amplifier. The present invention more particularlyrelates to a wide dynamic range transimpedance amplifier with automaticgain control and direct current (DC) cancellation control. The automaticgain control and direct current cancellation control are achieved in oneembodiment using variable impedance circuitry whose impedance iscontrolled by or related to the average photodiode current. The variableimpedance circuitry does not introduce significant open loop gain intothe low frequency DC/AGC cancellation feedback loop of thetransimpedance amplifier. In addition to automatic gain control anddirect current cancellation, the optical sensitivity of thetransimpedance amplifier is not reduced while the optical overload isincreased.

[0033] As the average photodiode current increases, the impedance of thevariable impedance circuitry decreases. The variable impedance circuitrycancels the DC component of the input signal and attenuates the ACcomponent of the input signal, thereby providing automatic gain controlwhile canceling the DC component of the input signal.

[0034]FIG. 1 illustrates an exemplary environment for implementingembodiments of the present invention. FIG. 1 illustrates a fiber opticreceiver that receives an optical signal (light) and converts theoptical signal to an electrical signal or data stream (usuallyrepresented as a voltage). The fiber optic receiver receives an opticalsignal over an optical fiber 102. A photo diode 104 or other opticaldevice that converts an optical signal to an electrical signal orcurrent (light to electrons conversion) receives the optical signal andgenerates an electrical signal 110 (current). The transimpedanceamplifier 120 amplifies the electrical signal 110 to produce theamplified electrical signal 112. The transimpedance amplifier 120 has awide dynamic range that is able to amplify signals with large powerwithout significantly diminishing the ability to amplify signals withlow power. The amplified electrical signal 112 is then translated by thetranslation module 108 and converted into an electrical digital signal114.

[0035]FIG. 2 illustrates a block diagram of an exemplary transimpedanceamplifier in accordance with the present invention. The transimpedanceamplifier 120 includes an input stage 122 that receives an electricalcurrent 110 from a photo diode or other device that converts an opticalsignal into the electrical voltage. An amplifier 124 amplifies theelectrical signal and helps reduce or prevent noise from being a factor.A buffer 126 is also optionally provided at the output of thetransimpedance amplifier 120. In one embodiment, the input stage 122 andthe amplifier 124 are referred to as an forward transimpedance circuit.It is understood by one of skill in the art that the input stage 122 andthe amplifier 124 can be implemented in different configurations.Exemplary configurations include, but are not limited to, a common baseconfiguration and a shunt feedback configuration. In addition, theamplifier 124 includes single ended amplification, differentialamplification, and the like or any combination thereof.

[0036] The feedback circuit 130 provides both automatic gain control anddirect current (DC) cancellation for the transimpedance amplifier 120.In the feedback circuit 130, a high frequency filter 132 is used todetect the DC component output by the amplifier 124. The DC component orlow frequency component of the output of the amplifier 124 is passed bythe high frequency filter 132 and is canceled by the variable impedancecircuitry 140. In another embodiment, the high frequency filter 132 maybe replaced with a peak detector or similar circuitry.

[0037] The variable impedance circuitry 140 also provides automatic gaincontrol for the transimpedance amplifier 120 because it is able toattenuate at least some of the AC content of the photodiode current whenthe impedance of the variable impedance circuitry 140 decreases. Inother words, the impedance of the variable impedance circuitry 140changes according to the average current of the photodiode. As theaverage current received from the photodiode or other source increases,the impedance of the variable impedance circuitry decreases. Because theimpedance of the variable impedance circuitry 140 decreases, thevariable impedance circuitry 140 absorbs or attenuates some of the ACcomponent. This provides automatic control of the transimpedance gain ofthe fiber optic receiver. When the average photodiode current is low,the impedance of the variable impedance circuitry 140 is relativelylarge and the AC component is not absorbed or attenuated, but isamplified at the input stage 122 and/or by the amplifier 124. Thus, theoptical overload of the transimpedance amplifier is increased withoutsimultaneously trading off the optical sensitivity of the transimpedanceamplifier.

[0038] In This is advantageous for the transimpedance amplifier 120because the range of signals that can be amplified without clipping,saturation, or other problems, is increased. Low power signals are alsoamplified by the transimpedance amplifier 120 as the AC component is notabsorbed or attenuated by the variable impedance circuitry 140, whileoptical signals with higher optical power are partially absorbed orattenuated by the variable impedance circuitry 140. The transimpedanceamplifier 120 can thereby successfully amplify a wide range of signals.

[0039]FIG. 3 illustrates one embodiment of a transimpedance amplifier120. The transimpedance amplifier 120 of FIG. 3 utilizes a common basetopology with a feedback circuit that provides both low frequency or DCcancellation and automatic gain control, as previously stated.Generally, the transimpedance amplifier includes an amplifier thatincludes one or more stages. The DC offset or component is sensed by thefeedback circuit and eliminated from the input signal. In the example ofFIG. 3, the transistors 200 and 202 are included in the input stage. Thecurrent from the photodiode is converted to a voltage by the transistor202. The voltage output by the transistor 200 serves as a referencevoltage in this embodiment. An output signal from the transistors 200and 202 is input to the transistors 206 and 208, which are arranged inan emitter follower configuration such that the voltage at the emitterssubstantially follows the voltage at the bases of the transistors 206and 208. The amplifier 210 amplifies the output of the emitter followers(206 and 208).

[0040] The DC or low frequency component of the output of the amplifier210 is passed by the low frequency operational amplifier 214, which isan example of a high frequency filter, and drives the base of the pnptransistor 204. Also, the DC or low frequency component can be sensed atthe output of the input stage or at the output of the output of theemitter follower transistors 206 and 208.

[0041]FIG. 3, the transistor 204 is a pnp transistor and the DCcomponent or low frequency component detected by the low frequencyoperational amplifier 214 drives the base of the pnp transistor 204. Theemitter of the pnp transistor 204 is also electrically connected withthe signal generated by the photodiode. As the average photodiodecurrent increases, the emitter impedance of the transistor 204decreases. This enables some of the AC component being processed by thetransistor 202 to be absorbed by the transistor 204 and permits thetransimpedance amplifier to amplify or transmit signals whose opticalpower is large. The transistor 204 is an example of the variableimpedance circuitry of FIG. 2.

[0042] Because the transimpedance amplifier shown FIG. 3 uses a pnptransistor instead of a npn transistor for the transistor 204 (Q₂), theAC content or component of the photodiode current will be absorbed orattenuated by the transistor 204 when the impedance seen at the emitterof the transistor 204 decreases. This is the case when the photodiodecurrent increases and the optical signal detected by the photodiode hasincreased power.

[0043] Furthermore, the pnp transistor 204 can be replaced with an npntransistor as long as the input signal from the photodiode is notconnected at the collector of the npn transistor. The input signal isconnected with the emitter of the npn transistor. Also the cathode ofthe photodiode connector is connected with the emitter of the npntransistor in this embodiment.

[0044] The variation of the input impedance at the emitter of thetransistor 204 with the average photodiode current provides an automaticcontrol of the transimpedance gain of the receiver with the averagephotodiode current. In contrast, when an npn transistor is utilizedinstead of a pnp transistor in the embodiment of FIG. 3 and thecollector of the npn transistor is connected to the anode of thephotodiode, the base o the output of 214 and the emitter to ground. TheAC component of the photodiode current is not attenuated because theimpedance seen at the collector of the npn transistor is not dependenton the average photo diode current. In addition, an npn transistorintroduces increased open loop gain in the frequency response as theaverage photodiode current increases. The pnp transistor 204 does notintroduce the gain that would otherwise be introduced by an npntransistor.

[0045]FIG. 4 illustrates another example of the variable impedancecircuitry 140. In this example, a npn transistor 302 is coupled with thepnp transistor 204. More specifically, the emitter of the transistor 302is connected to the emitter of the transistor 204. This permits thevariable impedance circuitry 140 to either source or sink the DC and ACcomponents of current and the photodiode can therefore be connected toeither a negative supply (or ground) or a positive supply. If thephotodiode 306 is connected to a negative supply or ground, then the npntransistor 302 has a variable impedance that depends on the averagecurrent of the photodiode 306. When the photodiode 304 is utilized, thenthe pnp transistor 204 has a variable impedance that is used forautomatic gain control through AC attenuation and DC cancellation.

[0046] With reference to FIGS. 2 and 3, the feedback circuit includes avery low gain-bandwidth op-amp (B(s)) driving the base of the transistor204 and/or the transistor 302 (Q₂₂). The feedback circuit senses the DCoffset at the output of the A₁ gain stage (210) or at the output of thetransistors 206 and 208, or other suitable location. Because the gainstage of the amplifier 210 is DC coupled to the input stage of thetransimpedance amplifier, any offset between the transistor 200 andtransistor 202 collector voltages resulting from a difference ofcollector current is compensated by the transistor 302 sourcing currentat the input of the transimpedance amplifier or the transistor 204sinking current at the input of the transimpedance amplifier.

[0047] As a result, the feedback loop or circuit 130 from the amplifieror gain stage to the input of the transimpedance amplifier removes theDC current or low frequency component of the photodiode signal.Therefore, the transconductance g_(m2) of the transistor 204 isproportional to the average photodiode current and hence the averagereceived optical power (assuming the internal offset generated by thetransimpedance amplifier is ignored).

[0048] For the transistor 302 (Q₂₂) or the transistor 204 (Q₂) thetransconductance is: $\begin{matrix}{{g_{m2} = \frac{I_{PD}}{V_{T}}},} & (1)\end{matrix}$

[0049] where I_(PD) is the average current of the photodiode or otheroptical device that converts an optical signal in to an electricalsignal such as current.

[0050] In the frequency range where the transimpedance amplifierjunction capacitances and the photodiode input capacitance can beignored, the closed loop transimpedance transfer function is given by:$\begin{matrix}{{\frac{V_{out}}{I_{in}} = \frac{\frac{g_{m1}}{g_{m1} + g_{m2}}{R_{C} \cdot A_{1}}}{1 + {\frac{g_{m1}}{g_{m1} + g_{m2}}{R_{C} \cdot A_{1} \cdot g_{m2} \cdot {B(s)}}}}}{or}} & (2) \\{\frac{V_{out}}{I_{in}} = \frac{A}{1 + {A\quad \beta}}} & (3)\end{matrix}$

[0051] where the forward gain$A = {\frac{g_{m1}}{g_{m1} + g_{m2}}{R_{C} \cdot A_{1}}}$

[0052] and the feedback gain β=g_(m2)·B(s). The transconductance of thecommon base input stage is g_(m1) and is set by the base voltage and theresistor R_(E) 201 in the emitter.

[0053] In this example, a low frequency dominant pole OP-AMP with a DCgain of B drives the base of the feedback transistors (pnp transistor204 (Q₂) and/or the npn transistor 302 (Q₂₂)). The feedback gain can bewritten: $\begin{matrix}{\beta = {g_{m2}\frac{B}{1 + \frac{s}{w_{0}}}}} & (4)\end{matrix}$

[0054] The transconductance of the transistors 200 and 202 depend on thevoltage V_(BASE) and the resistor R_(E) in series with their emitters.The bias of the input stage (I_(C(Q1)) and I_(C(Q0))) should beoptimized for bandwidth and noise. The bias of the input stage does notdepend on the average photodiode current and remains constant when theoptical power received at the photodiode changes.

[0055] The transimpedance amplifier 120 is examined below from theperspectives of low optical power and of high optical power. At lowoptical power, g_(m2)<<g_(m1) (or I_(PD)<<I_(C(Q1))). Therefore, thetransimpedance of the transimpedance amplifier transfer function can besimplified: $\begin{matrix}{\frac{V_{out}}{I_{in}} = \frac{R_{C} \cdot A_{1}}{1 + {R_{C} \cdot A_{1} \cdot g_{m2} \cdot {B(s)}}}} & (5)\end{matrix}$

[0056] In the signal frequency band at low optical power, thetransimpedance value of the transimpedance amplifier becomes:$\begin{matrix}{\frac{V_{out}}{I_{in}} = {R_{C} \cdot A_{1}}} & (6)\end{matrix}$

[0057] At high optical power, where g_(m1)<<g_(m2) (orI_(PD)>>I_(C(Q1))) $\begin{matrix}{\frac{V_{out}}{I_{in}} = \frac{\frac{g_{m1}}{g_{m2}} \cdot R_{C} \cdot A_{1}}{1 + {\frac{g_{m1}}{g_{m2}} \cdot R_{C} \cdot A_{1} \cdot g_{m2} \cdot {B(s)}}}} & (7)\end{matrix}$

[0058] In the signal frequency band at high optical power, thetransimpedance value becomes: $\begin{matrix}{\frac{V_{out}}{I_{in}} = {\frac{g_{m1}}{g_{m2}} \cdot R_{C} \cdot A_{1}}} & (8) \\{or} & \quad \\{\frac{V_{out}}{I_{in}} = {\frac{I_{C{({Q1})}}}{I_{PD}} \cdot R_{C} \cdot A_{1}}} & (9)\end{matrix}$

[0059] The low frequency feedback causes the closed loop gain frequencyresponse to have a low frequency cutoff given by: $\begin{matrix}{f_{{HPF} - {3d\quad B}} = {\frac{g_{m1}}{\left( {g_{m1} + g_{m2}} \right)} \cdot R_{C} \cdot A_{1} \cdot g_{m2} \cdot B \cdot f_{0}}} & (10)\end{matrix}$

[0060] where $f_{0} = \frac{w_{0}}{2\pi}$

[0061] and B is the DC gain of the op-amp.

[0062] At low optical power, g_(m2)<<g_(m1) (or I_(PD)<<I_(C(Q1))).Thus, equation (10) can be simplified and the low frequency cutoff isgiven by:

ƒ_(HPF-3dB) =R _(C) ·A ₁ ·g _(m2) ·B·ƒ ₀.  (11)

[0063] At high optical power, g_(m1)<<g_(m2) (or I_(PD)>>I_(C(Q1))) andequation (10) can be simplified and the low frequency cutoff is givenby:

ƒ_(HPF-3dB) =R _(C) ·A ₁ ·g _(m1) ·B·ƒ ₀.  (12)

[0064] The low cutoff frequency at high optical power is not dependenton the transistor 204 or on the transconductance of the transistor 204.The low cutoff frequency is controlled. The low cutoff frequencyrepresents the −3 dB low corner frequency in the frequency response ofthe transimpedance amplifier. The present invention places a limit orcontrols the low corner frequency at high optical power.

[0065] In contrast, a similar analysis applied to a circuit thatutilizes an npn transistor in place of the pnp transistor such that thecollector of the npn transistor is connected with the input signal orcurrent has a low frequency cutoff that is dependent on thetransconductance of the npn transistor. As the average photodiodecurrent increases, the npn transistor causes the transimpedanceamplifier to have a higher low frequency cutoff. One disadvantage isthat a transimpedance amplifier using an npn transistor in the place ofthe pnp transistor 204 is that the transimpedance amplifier does notfunction at lower frequencies for higher optical power or larger inputcurrents. The present invention, however, functions at lower frequenciesfor higher optical power or larger input currents. This permitsembodiments of the transimpedance amplifier to be integrated withexisting networks that operate at lower frequencies.

[0066]FIG. 5 is another embodiment of the automatic gain control lowfrequency feedback loop using a shunt feedback topology. The transistor502 can be replaced with the circuit illustrated in FIG. 4 toaccommodate both a negative and positive supply as previously discussed.

[0067] The same analysis can be made for the shunt-feedbacktransimpedance amplifier configurations that was made for the commonbase configuration of FIG. 3. Using nodal analysis on the small signalcircuit low frequency model of the transimpedance amplifier input stageshunt-feedback amplifier in FIG. 5, the transimpedance transfer functionof the transimpedance amplifier can be extracted. In the frequency rangewhere the transimpedance amplifier junction capacitances and thephotodiode input capacitance can be ignored, the closed looptransimpedance transfer function is given by: $\begin{matrix}{\frac{V_{out}}{I_{in}} = {{- \frac{\frac{g_{m1} \cdot R_{C} \cdot A_{1} \cdot R_{F}}{1 + {g_{m1} \cdot R_{C}} + {g_{m2} \cdot R_{F}}}}{1 + {\frac{g_{m1} \cdot R_{C} \cdot R_{F}}{1 + {g_{m1} \cdot R_{C}} + {g_{m2} \cdot R_{F}}} \cdot g_{m2} \cdot A_{1} \cdot {B(s)}}}}\text{or}}} & (13) \\{\frac{V_{out}}{I_{in}} = {- \frac{A}{1 + {A \cdot \beta}}}} & (14)\end{matrix}$

[0068] where$A = \frac{g_{m1} \cdot R_{C} \cdot A_{1} \cdot R_{f}}{1 + {g_{m1} \cdot R_{C}} + {g_{m2} \cdot R_{F}}}$

[0069] (forward gain) and β=g_(m2)·B(s) (feedback gain).

[0070] At low optical power, where R_(C)·g_(m1)>>R_(F)·g_(m2) orR_(F)·I_(PD)<<R_(C)·I_(C(Q1)), the transimpedance of the transimpedanceamplifier transfer function can be simplified as: $\begin{matrix}{\frac{V_{out}}{I_{in}} = \frac{R_{F} \cdot A_{1}}{1 + {R_{F} \cdot A_{1} \cdot g_{m2} \cdot {B(s)}}}} & (15)\end{matrix}$

[0071] In the signal frequency band, the transimpedance value becomes:$\begin{matrix}{\frac{V_{out}}{I_{in}} = {R_{F} \cdot A_{1}}} & (16)\end{matrix}$

[0072] At high optical power, where R_(C)·g_(m1)<<R_(F)·g_(m2) orR_(F)·I_(PD)>>R_(C)·I_(C(Q1)) $\begin{matrix}{\frac{V_{out}}{I_{in}} = \frac{\frac{g_{m1}}{g_{m2}} \cdot R_{C} \cdot A_{1}}{1 + {\frac{g_{m1}}{g_{m2}} \cdot R_{C} \cdot A_{1} \cdot g_{m2} \cdot {B(s)}}}} & (17)\end{matrix}$

[0073] In the signal frequency band, the transimpedance value becomes:$\begin{matrix}{\frac{V_{out}}{I_{in}} = {{\frac{g_{m1}}{g_{m2}} \cdot R_{C} \cdot A_{1}}\text{or}}} & (18) \\{\frac{V_{out}}{I_{in}} = {\frac{I_{C{({Q1})}}}{I_{PD}} \cdot R_{C} \cdot A_{1}}} & (19)\end{matrix}$

[0074] The low frequency feedback causes the closed loop gain frequencyresponse to have a low frequency cutoff given by: $\begin{matrix}{f_{{HPF} - {3{dB}}} = {\frac{g_{m1} \cdot R_{C} \cdot A_{1} \cdot R_{F}}{1 + {g_{m1} \cdot R_{C}} + {g_{m2} \cdot R_{F}}} \cdot g_{m2} \cdot B \cdot f_{0}}} & (20)\end{matrix}$

[0075] where $f_{0} = \frac{w_{0}}{2\pi}$

[0076] and B is the DC gain of the opamp.

[0077] At low optical power, R_(C)·g_(m1)>>R_(F)·g_(m2) orR_(F)·I_(PD)<<R_(C)·I_(C(Q1)). Therefore, equation (20) can besimplified and the low frequency cutoff is given by:

ƒ_(HPF-3dB) =R _(F) ·A ₁ ·g _(m2) ·B·ƒ ₀  (21)

[0078] At high optical power, R_(C)·g_(m1)<<R_(F)·g_(m2) orR_(F)·I_(PD)>>R_(C)·I_(C(Q1)). Therefore, equation (20) can besimplified and the low frequency cutoff is given by:

ƒ_(HPF-3dB) =R _(C) ·A ₁ ·g _(m1) ·B·ƒ  (22)

[0079] Again, the low frequency cutoff at high optical power is notdependent on the transconductance of the transistor 502.

[0080]FIG. 6 illustrates an alternative embodiment of the variableimpedance circuitry using field effect transistors instead of bipolarjunction transistors. The variable impedance circuitry can include manytypes of field effect transistors (MOSFETS, JFETS), BJT transistors, andthe like or any combination thereof. In another embodiment of thepresent invention, the photodiode that receives the optical signal is aphotodiode that also amplifies the optical signal or other device thatconverts the optical signal into an electrical signal or current.

[0081]FIG. 7 is a block diagram that plots the transimpedance of atransimpedance amplifier versus the average current of the photodiode.The plot 700 represents the transimpedance of an existing transimpedanceamplifier that utilizes, in one embodiment, an npn transistor for DCcancellation. For the plot 700, the collector of the npn transistor istypically connected with the input signal.

[0082] As the average photodiode current increases, the transimpedanceof existing transimpedance amplifiers as illustrated by the plot 700 isrelatively steady and does not drop until the current gets larger(generally because of the saturation of the input stage). The plot 702,on the other hand, illustrates that the transimpedance of thetransimpedance amplifier illustrated in FIG. 3 decreases as the averagephotodiode current increases. The plot 702 also illustrates that thetransimpedance increases as the average photodiode current decreases. Atthe point 704, for example, the transimpedance illustrated in the plot700 is much higher than the transimpedance of the plot 702.

[0083] The variable impedance circuitry of the present invention, whichincludes a pnp transistor in one embodiment, enables the transimpedanceamplifier to adjust to the transimpedance gain gradually as the averagephotodiode current increases or decreases.

[0084]FIG. 8 is a graph that plots the low corner frequency of thetransimpedance amplifier as it varies with the average photodiodecurrent. The plot 800 illustrates a plot of the low corner frequency ofexisting transimpedance amplifiers. The plot 802 illustrates a plot ofthe low corner frequency of a transimpedance amplifier in accordancewith the present invention. More particularly, the plot 802 representsthe low corner frequency of the transimpedance amplifier illustrated inFIG. 3.

[0085]FIG. 8 illustrates that as the average photodiode currentincreases, the low cutoff frequency of the plot 800 increases rapidlyand linearly. As previously described, this can ultimately result incircuit failure. In other words, the capability to transmit lower datarates at high input power optical sensitivity of the transimpedanceamplifier represented by the plot 800 is diminished. The diminishedtransmission capability results because the low cutoff frequency is notcontrolled and increases quickly as the input power of the opticalsignal or of the current from the optical device that converts theoptical signal to an input current increases.

[0086] In contrast, the low cutoff frequency of the transimpedanceamplifier represented by the plot 802 levels off or approaches an upperlimit as the average photodiode current increases. Because the increasein the low cutoff frequency in the plot 802 is substantially less thanthe increase illustrated by the plot 800, the transimpedance amplifierrepresented by the plot 802 can successfully accommodate a larger rangeof data rates. The optical sensitivity is improved and thetransimpedance amplifier can interact with legacy systems that mayoperate at lower frequencies.

[0087]FIG. 9 illustrates another embodiment of a transimpedanceamplifier with automatic gain control and DC cancellation circuitry.FIG. 9 is similar to FIG. 5 and also includes transistor 907.Transistors 902 and 907 are pnp transistors. The shunt-feedbacktransimpedance amplifier of FIG. 9 can be used to amplify signals havinga wide range of different optical powers. Using nodal analysis on thecircuit of FIG. 9, the transimpedance transfer function can beidentified. In FIG. 9, transistor 907 can be some factor N smaller inthe emitter size than transistor 902. Accordingly, since amplifier 906cancels DC signals:

I ₂ =I _(PD) +I _(RF1)  (23)

[0088] and $\begin{matrix}{I_{3} = \frac{I_{2}}{N}} & (24)\end{matrix}$

[0089] furthermore

I _(RF5) =I ₃  (25)

[0090] and

R _(F5) ·I _(RF5) =R _(F1) ·I _(RF1)  (26)

[0091] and thus

I _(RF5) =I _(RF1) =I ₃  (27)

[0092] From equations (23)-(27) it can be calculated that:$\begin{matrix}{I_{RF5} = {I_{RF1} = \frac{I_{PD}}{\left( {N - 1} \right)}}} & (28)\end{matrix}$

[0093] Additionally, since amplifier 906 cancels DC signals:$\begin{matrix}{V_{A} = {V_{B} = {V_{{BEQ}\quad 1} + {R_{F} \cdot \frac{I_{PD}}{\left( {N - 1} \right)}} + V_{{BEQ}\quad 5}}}} & (29)\end{matrix}$

[0094] and thus I_(Q1) (the current through Q₁) is given by:$\begin{matrix}{I_{Q\quad 1} = {\frac{V_{CC} - V_{A}}{R_{C}} = {\frac{V_{CC} - {2V_{BE}} - {R_{F} \cdot \frac{I_{PD}}{\left( {N - 1} \right)}}}{R_{C}} = {\frac{V_{CC} - {2V_{BE}}}{R_{C}} - {\left( \frac{R_{F}}{R_{C}} \right) \cdot \left( \frac{I_{PD}}{\left( {N - 1} \right)} \right)}}}}} & (30)\end{matrix}$

[0095] Thus, I_(Q1) decreases when the photodiode average currentincreases

[0096] The transconductance g_(m1) of transistor 901 (Q₁) is given by:$\begin{matrix}{g_{m\quad 1} = {\frac{I_{Q\quad 1}}{V_{T}} = {\left\lbrack {\frac{V_{CC} - {2V_{BE}}}{V_{T}} \cdot \left( \frac{1}{R_{C}} \right)} \right\rbrack - \left\lbrack {\left( \frac{R_{F}}{R_{C}} \right) \cdot \left( \frac{I_{PD}}{\left( {N - 1} \right)} \right) \cdot \left( \frac{1}{V_{T}} \right)} \right\rbrack}}} & (31)\end{matrix}$

[0097] At high optical power (and similar to equation 22), the lowfrequency cutoff is given by:

ƒ_(HPF-3dB) =R _(C) ·g _(m1) ·B·ƒ ₀  (32)

[0098] Similar to FIG. 5, the low frequency cutoff at high optical poweris not dependent on the transconductance of the transistor 902. Further,the low frequency cutoff can be adjusted by varying the value of N.Accordingly, the transconductance amplifier of FIG. 9 can providesufficient gain at a variety of frequencies.

[0099]FIG. 10 is an example plot of the gain of a transimpedanceamplifier versus frequency. As depicted in FIG. 10, the circuit of FIG.9 provides similar gain over a range of frequencies from frequency 1001to frequency 1002. In one embodiment, frequency 1001 could be set toaccommodate a data rate of 50 Mb/s and frequency 1002 could be set toaccommodate a data rate of 2.5 Gb/s (or an even higher data rate). Thus,similar gain is provided at data rates ranging from (and including) 50MB/s to 2.5 GB/s (or an even higher data rate).

[0100]FIG. 11 is an example of a number of plots of transimpedanceamplifier transimpedance gain versus photodiode average current. Plot1101 depicts an example of transimpedance amplifier transimpedance gainversus photodiode average current for a prior art circuit. Plot 1102depicts an example of transimpedance amplifier transimpedance gainversus photodiode average current for a circuit (e.g., the circuit ofFIG. 5) that uses a pnp transistor (e.g., transistor 502) for gaincontrol. Plot 1103 depicts an example of transimpedance amplifiertransimpedance gain versus photodiode average current for a circuit thatuses two pnp transistors, having an emitter size ratio M of 10, for gaincontrol. For example, plot 1103 could represent the circuit of FIG. 9where the emitter size of 902 divided by the emitter size of 907 equals10.

[0101] Plot 1104 depicts an example of transimpedance amplifiertransimpedance gain versus photodiode average current for a circuit thatuses two pnp transistors, having an emitter size ratio M of 5, for gaincontrol. For example, plot 1104 could represent the circuit of FIG. 9where the emitter size of 902 divided by the emitter size of 907 equals5. Plot 1105 depicts an example of transimpedance amplifiertransimpedance gain versus photodiode average current for a circuit thatuses two pnp transistors, having an emitter size ratio M of 2.5, forgain control. For example, plot 1105 could represent the circuit of FIG.9 where the emitter size of 902 divided by the emitter size of 907equals 2.5.

[0102] As depicted in FIG. 11, as photodiode average current increases,the transimpedance gain for circuits that utilize two pnp transistorsfor gain control (e.g., the circuit of FIG. 9) decreases more quicklythan the transimpedance gain for prior art circuits and for circuitsthat utilize one pnp transistor for gain control (e.g., the circuit ofFIG. 5). For example, plots 1103, 1104, and 1105 decrease more quicklythan plot 1102. Further, as photodiode average current increases, thetransimpedance gain for circuits having lower emitter size ratiosdecreases more quickly than the transimpedance gain for circuits havinggreater emitter size ratios. For example, plot 1105 (M=2.5) decreasesmore quickly than both plot 1103 (M=10) and plot 1104 (M=5).

[0103]FIG. 12 is an example of a number of plots of transimpedanceamplifier low frequency cut-off versus photodiode average current. Plot1201 depicts an example of transimpedance amplifier low frequencycut-off versus photodiode average current for a prior art circuit. Plot1202 depicts an example of transimpedance amplifier low frequencycut-off versus photodiode average current for a circuit (e.g., thecircuit of FIG. 5) that uses a pnp transistor (e.g., transistor 502) forgain control. Plot 1203 depicts an example of transimpedance amplifierlow frequency cut-off versus photodiode average current for a circuitthat uses two pnp transistors, having an emitter size ratio M of 10, forgain control. For example, plot 1203 could represent the circuit of FIG.9 where the emitter size of 902 divided by the emitter size of 907equals 10.

[0104] Plot 1204 depicts an example of transimpedance amplifier lowfrequency cut-off versus photodiode average current for a circuit thatuses two pnp transistors, having an emitter size ratio M of 5, for gaincontrol. For example, plot 1204 could represent the circuit of FIG. 9where the emitter size of 902 divided by the emitter size of 907 equals5. Plot 1205 depicts an example of transimpedance amplifier lowfrequency cut-off versus photodiode average current for a circuit thatuses two pnp transistors, having an emitter size ratio M of 2.5, forgain control. For example, plot 1203 could represent the circuit of FIG.9 where the emitter size of 902 divided by the emitter size of 907equals 2.5.

[0105] As depicted in FIG. 12, as photodiode average current increases,the low frequency cut-off for circuits that utilize two pnp transistorsfor gain control (e.g., the circuit of FIG. 9) is more controlled thanthe low frequency cut-off for prior art circuits and for circuits thatutilize one pnp transistor for gain control (e.g., the circuit of FIG.5). For example, plots 1203, 1204, and 1205 begin transition lower oncephotodiode average current reaches a threshold value. Further, asphotodiode average current increases, the low frequency cut-off forcircuits having lower emitter size ratios transitions to decreasing morequickly than for circuits having greater emitter size ratios. Forexample, plot 1205 (M=2.5) transitions to decreasing more quickly thanboth plot 1203 (M=10) and plot 1204 (M=5).

[0106] The present invention may be embodied in other specific formswithout departing from its spirit or essential characteristics. Thedescribed embodiments are to be considered in all respects only asillustrative and not restrictive. The scope of the invention is,therefore, indicated by the appended claims rather than by the foregoingdescription. All changes which come within the meaning and range ofequivalency of the claims are to be embraced within their scope.

What is claimed is:
 1. A transimpedance amplifier circuit with anamplifier input, the transimpedance amplifier circuit having acontrolled low cutoff frequency as average input current to theamplifier input increases, the transimpedance amplifier circuitcomprising: an optical device having an optical device input terminaland an optical device output terminal, the optical device for receivingan optical signal at the optical device input terminal, converting theoptical signal to an input current, and providing the input current atthe optical device output terminal; a forward transimpedance circuitconnected to the optical device output terminal, the forwardtransimpedance circuit for receiving the input current and generating anoutput signal based on the input current; a feedback circuit thatincludes: a first circuit that for detecting a low frequency componentof the output signal; and a second circuit that is driven by the lowfrequency component of the output signal and is connected to the forwardtransimpedance circuit such that the impedance of the second circuitpresented at the amplifier input decreases as the output signalincreases, the second circuit including: a first pnp transistor having afirst base terminal and a first emitter terminal, wherein the firstemitter terminal is connected to the amplifier input; and a second pnptransistor having a second base terminal, the second base terminal beingconnected to the first base terminal, the second pnp transistor havingan emitter size that is some factor smaller than an emitter size of thefirst pnp transistor.
 2. A transimpedance amplifier circuit as recitedin claim 1, wherein an impedance seen at the first emitter terminal isdependent on the average current of the input current and wherein thelow cutoff frequency does not increase linearly as the input currentincreases.
 3. A transimpedance amplifier circuit as recited in claim 1,wherein the second circuit has variable impedance such that increasingan optical overload of the transimpedance amplifier does not diminish anoptical sensitivity of the transimpedance amplifier.
 4. A transimpedanceamplifier circuit as recited in claim 1, wherein the first circuit andthe second circuit shunt a DC component of the input current such that aDC component of the output signal is significantly reduced.
 5. Atransimpedance amplifier circuit as recited in claim 1, wherein thefirst circuit includes a low frequency operational amplifier.
 6. Atransimpedance amplifier with an amplifier input, the transimpedanceamplifier having a controlled low cutoff frequency as average input tothe transimpedance amplifier increases, the transimpedance amplifiercomprising: an input stage for receiving an input current signalprovided at the output terminal of an optical device, the input stagegenerating an output voltage based on the received input current; a gainstage for amplifying the output voltage to generate an amplified signal;and a feedback circuit that includes: a low frequency circuit fordetecting a low frequency component of the amplified signal such thatthe low frequency component can be removed from the amplified signal;and variable impedance circuitry, wherein an impedance of the variableimpedance circuitry is dependent on an average current of the inputcurrent signal such that the impedance decreases as the average currentincreases and wherein a low cutoff frequency of the transimpedanceamplifier decreases when the average current increases to greater than aspecified threshold, the variable impedance circuitry including: a firstpnp transistor having a first base terminal and a first emitterterminal, wherein the emitter terminal is connected to the amplifierinput, the first pnp transistor have a first emitter size; and a secondpnp transistor having a second base terminal, the second base terminalbeing connected to the first base terminal, the second pnp transistorhaving a second emitter size that is some factor smaller than the firstemitter size.
 7. A transimpedance amplifier as recited in claim 6,wherein the input stage is in a shunt feedback configuration and whereinthe gain stage is an amplifier.
 8. A transimpedance amplifier as recitedin claim 6, wherein the low frequency circuit further comprises a lowfrequency operational amplifier.
 9. A transimpedance amplifier asrecited in claim 6, wherein the low frequency circuit detects andreduces the low frequency component at the input stage by shunting thelow frequency component of the input current signal.
 10. Atransimpedance amplifier as recited in claim 6, wherein the first pnptransistor that has a transconductance that does not affect the lowcutoff frequency of the transimpedance amplifier as the input currentsignal increases.
 11. In a system that receives input currents ofdifferent magnitudes, the system including a forward transimpedancecircuit, a low frequency detection circuit, a variable impedancecircuit, a first pnp transistor, and a second pnp transistor, the firstpnp transistor having a first base terminal and a first emitter size,the second pnp transistor having a second base terminal, the second baseterminal being connected to the first base terminal, the second pnptransistor having a second emitter size that is some factor smaller thanthe first emitter size, a method for controlling a low cutoff frequencyas average input current to the system increases, the method comprising:the forward transimpedance circuit receiving an input current from anoptical device; the forward transimpedance circuit generating an outputsignal based on the input current; the low frequency detection circuitdetecting a low frequency component of the output signal; utilizing thelow frequency component to determine the impedance of the variableimpedance circuit such that the impedance of the of the variableimpedance circuit decreases as the average input current increases;utilizing the first and second pnp transistors to control the low cutofffrequency such that the low cutoff frequency transitions to decreasingwhen the magnitude of average input current reaches a specifiedthreshold.